Bias generator for a four transistor load less memory cell

ABSTRACT

The present invention is a current-mirror-based bias generator for a load less four transistor SRAM as well as associated methods of controlling or modifying the current conducted by the access transistors of such an SRAM. The present invention may be thought of as an adjustable temperature coefficient, bias generator that references, via a current mirror, a reference bank of SRAM cells. The bank of reference cells provides an indication of the necessary conduction characteristics (e.g., gate to source voltage) of the access transistors under various conditions. By applying a bias voltage to the word line the desired current is sourced from the digit line. The bank of reference SRAM cells inherently compensates for process variations. The adjustable temperature coefficient bias generator allows the current sourced by the digit lines to vary greatly as a result of temperature variations. Thus, the present invention compensates for both process variations and temperature variations.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates generally to static-random-access-memory (SRAM)devices and, more particularly, to SRAM's utilizing a four transistordesign.

2. Description of the Background

To meet customer demand for smaller and more power efficient integratedcircuits (ICs), manufacturers are designing newer ICs that operate withlower supply voltages and that include smaller internal subcircuits suchas memory cells. Many ICs, such as memory circuits or other circuitssuch as microprocessors that include onboard memory, include arrays ofSRAM cells for data storage. SRAM cells are popular because they operateat a higher speed than dynamic-random-access-memory (DRAM) cells, whichmust be periodically refreshed.

FIG. 1 is a circuit diagram of a conventional 6-transistor (6-T) SRAMcell 10, which can operate at a relatively low supply voltage, forexample 1.5V-3.3V, but which is relatively large. A pair of NMOS accesstransistors 12 and 14 allow complementary bit values D and {overscore(D)} on digit lines 16 and 18, respectively, to be read from and to bewritten to a storage circuit 20 of the cell 10. The storage circuit 20includes NMOS pull-down transistors 22 and 26, which are coupled in apositive-feedback configuration with PMOS pull-up transistors 24 and 28,respectively. Nodes A and B are the complementary inputs/outputs of thestorage circuit 20, and the respective complementary logic values atthese nodes represent the state of the cell 10. For example, when thenode A is at logic 1 and the node B is at logic 0, then the cell 10 isstoring a logic 1. Conversely, when the node A is at logic 0 and thenode B is at logic 1, then the cell 10 is storing a logic 0. Thus, thecell 10 is bistable, i.e., the cell 10 can have one of two stablestates, logic 1 or logic 0.

In operation during a read of the cell 10, a word-line WL, which iscoupled to the gates of the transistors 12 and 14, is driven to avoltage approximately equal to Vcc to activate the transistors 12 and14. For example purposes, assume that Vcc=logic 1=5V and Vss=logic 0=0V,and that at the beginning of the read, the cell 10 is storing a logic 0such that the voltage level at the node A is 0V and the voltage level atthe node B is 5V. Also, assume that before the read cycle, the digitlines 16 and 18 are equilibrated to approximately Vcc-Vt. Therefore, theNMOS transistor 12 couples the node A to the digit line 16, and the NMOStransistor 14 couples the node B to the digit line 18. For example,assume that the threshold voltages of the transistors 12 and 14 are both1V, then the transistor 14 couples a maximum of 4V from the digit line18 to the node B. The transistor 12, however, couples the digit line 16to the node A, which pulls down the voltage on the digit line 16 enough(for example, 100-500 millivolts) to cause a sense amp (not shown)coupled to the lines 16 and 18 to read the cell 10 as storing a logic 0.

In operation during a write, for example, of a logic 1 to the cell 10,and making the same assumptions as discussed above for the read, thetransistors 12 and 14 are activated as discussed above, and logic 1 isdriven onto the digit line 16 and a logic 0 is driven onto the digitline 18. Thus, the transistor 12 couples 4V (the 5V on the digit line 16minus the 1V threshold of the transistor 12) to the node A, and thetransistor 14 couples 0V from the digit line 18 to the node B. The lowvoltage on the node B turns off the NMOS transistor 26, and turns on thePMOS transistor 28. Thus, the inactive NMOS transistor 26 allows thePMOS transistor 28 to pull the node A up to 5V. This high voltage on thenode A turns on the NMOS transistor 22 and turns off the PMOS transistor24, thus allowing the NMOS transistor 22 to reinforce the logic 0 on thenode B. Likewise, if the voltage written to the node B is 4V and thatwritten to the node A is 0V, the positive-feedback configuration ensuresthat the cell 10 will store a logic 0.

Because the PMOS transistors 24 and 28 have low on resistances(typically on the order of a few kilohms), they can pull the respectivenodes A and B virtually all the way up to Vcc often in less than 10nanoseconds (ns), and thus render the cell 10 relatively stable andallow the cell 10 to operate at a low supply voltage as discussed above.But unfortunately, the transistors 26 and 28 cause the cell 10 to beapproximately 30%-40% larger than a 4-transistor (4-T) SRAM cell, whichis discussed next.

FIG. 2 is a circuit diagram of a conventional 4-T SRAM cell 30, whereelements common to FIGS. 1 and 2 are referenced with like numerals. Amajor difference between the 6-T cell 10 and the 4-T cell 30 is that thePMOS pull-up transistors 24 and 28 of the 6-T cell 10 are replaced withconventional passive loads 32 and 34, respectively. For example, theloads 32 and 34 are often polysilicon resistors. Otherwise thetopologies of the 6-T cell 10 and the 4-T cell 30 are the same.Furthermore, the 4-T cell 30 operates similarly to the 6-T cell 10.Because the loads 32 and 34 are usually built in another level above theaccess transistors 12 and 14 and the NMOS pull-down transistors 22 and26, the 4-T cell 30 usually occupies much less area than the 6-T cell10.

Additional, complex steps are required to form the load elements 32 and34 such that 4-T cells present the usual complexity versus costtradeoff. The high resistance values of the loads 32 and 34 cansubstantially lower the stability margin of the cell 30 as compared withthe cell 10. Thus, under certain conditions, the cell 30 caninadvertently become monostable or read unstable instead of bistable.Also, the cell 30 consumes more power than the cell 20 because there isalways current flowing from Vcc to Vss through either the load 32 andthe NMOS transistor 26 or the load 34 and the NMOS transistor 22. Incontrast, current flow from Vcc to Vss in the cell 20 is always blockedby one of the NMOS/PMOS transistor pairs 22/24 and 26/28. Efforts toeliminate load elements 32 and 34 have lead to the development of aload-less four transistor SRAM cell as shown in FIG. 3.

FIG. 3 is a circuit diagram of a conventional load-less 4-T SRAM cell,where elements common to FIGS. 2 and 3 are referenced with likenumerals. The difference between the load-less cell 36 and the cell 30is the elimination of load elements 32 and 34 and the replacement ofNMOS transistors 12 and 14 with PMOS transistors 38 and 40,respectively.

With the load-less 4-T SRAM cell of FIG. 3, like all SRAM cells, leakagecurrents and/or subthreshold currents are generated by transistors 22and 26 when in the off state, and one will always be in the off state.To prevent the cell 36 from spontaneously changing state, thetransistors 38 and 40 must source sufficient load current from the digitlines 16 and 18, respectively, to offset the leakage and subthresholdcurrents. The needed load current can vary over many orders of magnitudedue to temperature and process variations. However, the load currentcannot be too large because the cumulative (along the digit lines 16 and18) load current needs to be significantly less than the cell currentfor proper noise margin for proper operation of the sense amps.

Wide temperature variations resulting from cold-data retention testingand burn-in testing are also causes of wide variations in leakage andsubthreshold currents, thereby causing wide variations in the loadcurrent that must be sourced by transistors 38 and 40. Such testing,coupled with normal process variations, sense amp margin requirements,as well as yield requirements (e.g., read/write stability requirements,power consumption requirements, etc.) have made the manufacturing ofload-less 4-T SRAM's a difficult matter.

SUMMARY OF THE PRESENT INVENTION

The present invention is directed generally to a bias generator used inconjunction with one of the word line or digit line to set the desiredlevel of load current as a function of temperature (or test beingperformed) to satisfy the simultaneous constraints of yield, sense ampmargin, and load current even during cold-data retention testing orburn-in.

The present invention is also directed to a method of modifying thelevel of current conducted by the access transistors of a load-less,four transistor memory cell when the access transistors are in an offstate. The method is comprised of the step of generating a temperaturedependent bias voltage and connecting that bias voltage to the gateterminals of the access transistors.

The present invention is also directed to a current-mirror-based biasgenerator for a load-less four transistor SRAM as well as associatedmethods of controlling or modifying the current conducted by the accesstransistors of such an SRAM. The present invention may be thought of asan adjustable temperature coefficient, bias generator that references,via a current mirror, a reference bank of SRAM cells. The bank ofreference cells provides an indication of the necessary conductioncharacteristics (e.g., gate to source voltage) of the access transistorsunder various conditions. By applying a bias voltage to the word linethe desired current is sourced from the digit line. The bank ofreference SRAM cells inherently compensates for process variations. Theadjustable temperature coefficient bias generator allows the currentsourced by the digit lines to vary greatly as a result of temperaturevariations. Thus, the present invention compensates for both processvariations and temperature variations. Those benefits, and others, willbecome apparent from the description of the preferred embodimenthereinbelow.

BRIEF DESCRIPTION OF THE DRAWINGS

For the present invention to be easily understood and readily practiced,the invention will now be described for purposes of illustration and notlimitation, in conjunction with the following figures wherein:

FIG. 1 is a circuit diagram of a conventional 6-T SRAM cell;

FIG. 2 is a circuit diagram of a conventional 4-T SRAM cell;

FIG. 3 is a circuit diagram of a conventional load-less 4-T SRAM cell;

FIG. 4 is a circuit diagram of a load-less 4-T SRAM cell in conjunctionwith a bias generator constructed according to the teachings of thepresent invention;

FIG. 5 is a block diagram of a portion of an array of 4-T SRAM cellsincorporating the bias generator of the present invention;

FIG. 6 is a block diagram of a load-less 4-T SRAM incorporating the biasgenerator of the present invention;

FIG. 7 is a block diagram of a computer system that includes the 4-TSRAM of FIG. 6;

FIG. 8 illustrates another embodiment of the present invention; and

FIG. 9 is a block diagram illustrating how the bias voltage on theglobal bus can be forced to any value.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 4 is a circuit diagram of a load-less 4 transistor SRAM cell 36 inconjunction with a bias generator 42 constructed according to theteachings of the present invention. The cell 36 illustrated in FIG. 4 isidentical to the cell 36 illustrated in FIG. 3. The bias generator 42 iscomprised of a bank of transistors 44 connected in parallel with eachother and connected in series with a temperature dependent constantcurrent source 46. The bank of transistors 44 is also connected inseries to a voltage source through a trimmable resistor 47. The bank oftransistors 44 is fabricated at the same time, and in the same manner,as access transistors 38 and 40. In that manner, the voltage drop fromthe gate terminal to the source terminal of each of the transistors 44should be substantially the same as the gate to source terminal drop ofaccess transistors 38 and 40. Thus, the voltage drop across the gate andsource terminals of each of the transistors 44 is representative of thevoltage drop across the gate and source terminals of the transistors 38and 40. As an alternative to separately fabricating the bank oftransistors 44, a bank of cells 36 could carry additional wiring so thatthe gate to source voltage of the access transistors 38 and 40 can besensed.

Returning to the bias generator 42, the current source 46 may beconstructed using any known techniques which provide a temperaturedependent constant current source. The constant current source willproduce one value of current under, for example, cold data-retentiontest conditions, and another value of current under burn-in testconditions. Thus, for each value of current produced by the temperaturedependent constant current source 46, a different voltage drop acrossthe gate and source terminals of the transistors 44 is produced. Thatvoltage drop is averaged and sensed by an operational amplifier 48.Although the bias generator 42 would operate if only one transistor forthe bank of transistors 44 was provided, by providing a plurality oftransistors within bank 44, a voltage drop which is more representativeof the voltage drop experienced in the cells is produced. The voltagedrop sensed by the operational amplifier 48 may then be applied to theword line which, as seen in the figure, is connected to the gateterminals of the access transistors 38 and 40. Thus, when the cell 36 isin the off-state, i.e., transistors 38 and 40 are nonconductive, thebias voltage applied by the operational amplifier 48 may be used tocontrol the conduction characteristics of the access transistors 38 and40 so as to enable the transistors 38 and 40 to source current from thedigit lines 16 and 18. Because the bias voltage is directly related tothe current which is produced by the constant current source 46, and thecurrent is temperature dependent, the bias voltage is also temperaturedependent. Thus, the conduction characteristics of the accesstransistors 38 and 40 are controlled according to the temperature suchthat the current required by the cell 36, for a given temperature, maybe properly sourced.

The temperature dependent constant current source may receive inputsfrom a programmable device 45. The programmable device 45 may containlaser trimmable devices, fuses, or antifuses, which allow manipulationof a value adjust signal (VA) and a temperature coefficient adjustsignal (TCA) to provide some degree of control over the bias voltagepost fabrication.

FIG. 5 is a block diagram of a portion of an array 50 of four transistorSRAM cells 36 incorporating the bias generator 42 of the presentinvention. In FIG. 5, a plurality of digit lines D, {overscore (D)}, anda plurality of word lines WLI-WL4 are used to interconnect individualmemory cells 36. The bias generator 42, constructed as shown in FIG. 4,globally provides the bias voltage to the array via global bus 54. Thebias generator 42 may be coupled to each of the word lines WL1-WL4through a transistor pair 52. Each transistor pair 52 is comprised of aPMOS and an NMOS transistor. The PMOS transistor may be connectedbetween the bias generator 42 and a word line, e.g., WL1. The NMOStransistor may be connected between the word line, e.g. WL1, and ground.Each transistor is responsive to a word line select signal, e.g. SelWL1.

In operation, only one word line will be active at a time. For wordlines not selected, the NMOS transistor of the transistor pair 52 willbe off while the PMOS transistor will be on thereby coupling the biasvoltage to each of the non-selected word lines. When a word line isselected, e.g., WL1, the word line select signal, e.g., Sel WL1, willcause the transistors to change state. Specifically, the NMOS transistorwill turn on connecting the word line to ground thereby rendering theword line active while the PMOS transistor will turn off thereby endingthe application of the bias voltage to the active word line.

To provide a particular voltage for a test mode, a voltage source 56 maybe coupled to the global bus 54 through a transistor 58. The voltagesource may be capable of outputting different voltages depending uponone or more control signals 60. Upon assertion of the signal {overscore(Tm)}, the bias generator 42 is disabled and the output of the voltagesource 56 is applied to the global bus 54. Voltage source 56 may includea constant current source as well as a laser trimmable device, fuses, orantifuses as discussed above for the purpose of giving the manufacturersome degree of control over the voltage(s) produced by the voltagesource 56 post fabrication.

Another way to implement the functionality described in the previousparagraph is through the use of more than one constant current source inthe bias generator 42 as shown in FIG. 8. For example, a second constantcurrent source 46′ could be operatively connected through a switch 66 tothe remainder of the circuit for producing a voltage input to op amp 48.The constant current source 46′ is responsive to a particular test modeinstead of being responsive to the temperature.

Another embodiment of the present invention is illustrated in FIG. 9. InFIG. 9, a pad 62 is connected to the global bus 54 through a transistor64. Whenever the signal {overscore (Tm)}-force is asserted, the biasgenerator 42 is disabled and the voltage available at the pad 62 isplaced on the global bus 54. Thus, the voltage on global bus 54 may beforced to any value. When the signal {overscore (Tm)}-measure isasserted, the voltage on bus 54 can be measured at pad 62. Thisfunctionality is useful for characterization purposes as well as yieldand reliability screening.

FIG. 6 is a block diagram of a memory circuit 70 which can include cells36 and the bias generator 42 as previously described. In one embodiment,the memory circuit 70 may be a synchronous SRAM.

The memory circuit 70 includes an address register 72, which receives anaddress from an ADDRESS bus (not shown). A control logic circuit 74receives a clock (CLK) signal, and receives enable and write signals ona COMMAND bus (not shown), and communicates with the other circuits ofthe memory circuit 70. A burst counter 75 causes the memory circuit 70to operate in a burst address mode in response to a MODE signal.

During a write cycle, write driver circuitry 76 writes date to a memoryarray 78. The array 78 is the component of the memory circuit 70 thatcan include the cells 36 and bias generator 42. The array 78 alsoincludes an address decoder 80 for decoding the address from the addressregister 72. Alternately, the address decoder 80 may be separate fromthe array 78.

During a read cycle, sense amplifiers 82 amplify and provide the dataread from the array 78 to a data input/output (I/O) circuit 84. The I/Ocircuit 84 includes output circuits 86, which provide data from thesense amplifiers 82 to a DATA bus (not shown) during a read cycle. TheI/O circuit 84 also includes input circuits 88, which provide data fromthe DATA bus to the write drivers 76 during a write cycle. The input andoutput circuits 88 and 86, respectively, may include conventionalregisters and buffers. Furthermore, the combination of the write drivercircuitry 76 and the sense amplifiers 82 can be referred to asread/write circuitry. The various components shown in FIG. 6, with theexception of the array 78, constitute a plurality of components forreading information out of, and writing information into, the array 80.

FIG. 7 is a block diagram of an electronic system 90, such as a computersystem, that incorporates the memory circuit 70 of FIG. 6. The system 90includes computer circuitry 92 for performing computer functions, suchas executing software to perform desired calculations and tasks. Thecircuitry 92 typically includes a processor 94 and the memory circuit70, which is coupled to the processor 94. One or more input devices 96,such as a keyboard or a mouse, are coupled to the computer circuitry 92and allow an operator (not shown) to manually input data thereto. One ormore output devices 98 are coupled to the computer circuitry 92 toprovide to the operator data generated by the computer circuitry 92.Examples of such output devices 98 include a printer and a video displayunit. One or more data-storage devices 100 are coupled to the computercircuitry 92 to store data on or retrieve data form external storagemedia (not shown). Examples of the storage devices 100 and thecorresponding storage media include drives that accept hard and floppydisks, tape cassettes, and compact disk read-only memories (CD-ROMs).Typically, the computer circuitry 92 includes address, data, and commandbuses and a clock line that are respectively coupled to the ADDRESS,DATA and COMMAND buses, and the CLK line of the memory circuit 70.

The present invention is also directed to a method of controlling theload current in a load-less four transistor memory cell. The method iscomprised of the step of providing a temperature dependent bias voltageto one of the word line or the digit line. The providing step may becomprised of the steps of generating a temperature dependent constantcurrent, generating a voltage drop across two terminals of a transistorrepresentative of the transistors in the memory cell with thetemperature dependent constant current, and sensing the voltage drop toproduce the bias voltage. The voltage drop may be generated across aplurality of transistors to provide an average value for the voltagedrop. By connecting or applying the bias voltage to one of the word lineor digit line, the conduction of the access transistors of the memorycell may be controlled. However, because of the different function whichthe digit line performs in the context of a memory cell, it isconsidered preferable to apply the bias voltage to the word line. Thepresent invention is also directed to a method of regulating a voltagedifference between the word line and the digit line in a load-less fourtransistor memory cell by applying a temperature dependent bias voltageto one of the word line or the digit line.

While the present invention has been described in conjunction withpreferred embodiments thereof, those of ordinary skill in the art willrecognize that many modifications and variations are possible. Forexample, the type of transistors used to construct the cell may bevaried such that the terminals in issue need not be the gate and sourceterminals. As previously mentioned, the same result can be achieved byvarying the voltage on the digit line or, alternatively, controlling thevoltage differential between the word line and the digit line. Theforegoing disclosure and the following claims are intended to encompassall such modifications and variations.

What is claimed is:
 1. A voltage generator for providing a bias voltageto a word line, comprising: a constant current source for producingdifferent levels of current in response to different conditions; atleast one transistor connected in series with said constant currentsource; and an amplifier responsive to a voltage across two terminals ofsaid transistor for producing a bias voltage.
 2. The voltage generatorof claim 1 wherein said constant current source is responsive todifferent temperature conditions.
 3. The voltage generator of claim 1additionally comprising a laser trimmable device connected with saidconstant current source.
 4. The voltage generator of claim 1additionally comprising a plurality of fuses connected with saidconstant current source.
 5. The voltage generator of claim 1additionally comprising a plurality of antifuses connected with saidconstant current source.
 6. A temperature dependent voltage generator,comprising: a temperature dependent constant current source forproducing different levels of current in response to differenttemperatures; at least one transistor connected in series with saidconstant current source; and an amplifier responsive to a voltage acrosscertain terminals of said transistor for providing a voltage.
 7. Thevoltage generator of claim 6 wherein said amplifier is responsive to avoltage across a gate terminal and a source terminal.
 8. The voltagegenerator of claim 6 additionally comprising means connected with saidconstant current source for controlling the voltage post fabrication. 9.A temperature compensated bias generator for use with an array ofloadless four transistor memory cells interconnected by bit lines andword lines, said bias generator comprising: a temperature dependentconstant current source; a plurality of transistors connected inparallel with one another and in series with said constant currentsource; and an amplifier responsive to an average voltage across twoterminals of said plurality of transistors for providing a bias voltage.10. The bias generator of claim 9 wherein said average voltage acrosstwo terminals includes the average voltage across a gate terminal and asource terminal of each transistor in said plurality of transistors.